Reconfigurable load modulation amplifier

ABSTRACT

A reconfigurable load modulation amplifier having a carrier amplifier and a peak amplifier that are coupled in parallel is disclosed. The peak amplifier provides additional power amplification when the carrier amplifier is driven into saturation. A quadrature coupler coupled between the carrier amplifier and the peak amplifier is configured to combine power from both the carrier amplifier and the peak amplifier for output through an output load terminal. The reconfigurable load modulation amplifier further includes control circuitry coupled to an isolation port of the quadrature coupler and configured to provide adjustable impedance at the isolation port of the quadrature coupler. As such, impedance at the isolation port of the quadrature coupler is tunable such that at least a carrier or peak amplifier is presented with a quadrature coupler load impedance that ranges from around about half an output load termination impedance to around about twice the output load termination impedance.

RELATED APPLICATIONS

This application claims the benefit of U.S. provisional patentapplication No. 61/884,571, filed Sep. 30, 2013, and U.S. provisionalpatent application No. 61/892,683, filed Oct. 18, 2013, the disclosuresof which are hereby incorporated herein by reference in theirentireties.

FIELD OF THE DISCLOSURE

The present disclosure pertains to amplifiers and in particular to loadmodulation amplifiers having a carrier amplifier and a peak amplifierthat are coupled in parallel.

BACKGROUND

The quest for high efficiency over a large peak to average power ratio(PAPR) and the ability to cover a wide bandwidth (BW) of operation (>160MHz and typically >5% of carrier frequency) is desired for future highdata rate wireless transmissions. Envelope Tracking (ET) has beendemonstrated to provide high efficiency across >10 dB of PAPR for 20 MHzlong term evolution (LTE) mobile applications. However, this approach isBW challenged and is strongly dependent on fundamental semiconductortechnology trades. Bandwidths of up to 30 MHz have been demonstrated formobile applications. Wireless Fidelity (Wi-Fi) BW requirements for 160MHz 802.11 ac are challenging without costly semiconductor technologydisruptions.

Doherty amplifiers can be configured to provide high efficiency over a 6dB OPBO (output power back off or PAPR range). In particular, Dohertyamplifiers may be configured to achieve larger PAPRs of 9 dB and 12 dBor more using N-way and/or asymmetric Doherty approaches. However, theseare typically limited to operating BW<5% due to the narrow-band quarterwave impedance inverters used. Doherty amplifiers also rely on DigitalPre-Distortion (DPD) to correct for phase and amplitude distortionresulting from the Doherty operation. For base stations, DPD has shownoperation up to 60 MHz with capability in the hundreds of MHz andexceeding 1 GHz for future point to point radio systems. However,Doherty amplifiers are not able to keep up with the BW capability ofDPD.

Thus, what is needed is a new amplifier paradigm that is reconfigurableto provide the advantageous characteristics of Doherty amplifieroperation while providing the BW capability of DPD. Moreover, anassociated need is a load modulation type amplifier that is configuredfor improved OPBO efficiency by reducing impedance at lower powerlevels. It is important to note that this need is contrary toconventional load modulation amplifier approaches.

SUMMARY

A reconfigurable load modulation amplifier having a carrier amplifierand a peak amplifier that are coupled in parallel is disclosed. The peakamplifier provides additional power amplification when the carrieramplifier is driven into saturation. A quadrature coupler that iscoupled between the carrier amplifier and the peak amplifier isconfigured to combine power from both the carrier amplifier and the peakamplifier for output through an output load terminal.

In at least some of the embodiments, the carrier amplifier is coupled tothe output load terminal through zero phase shift ports of thequadrature coupler. Moreover, at least some embodiments include controlcircuitry at an isolation port of the quadrature coupler. The controlcircuitry is configured to provide adjustable impedance at the isolationport of the quadrature coupler. Further still, in at least someembodiments, the reconfigurable load modulation amplifier includesvoltage standing wave ratio (VSWR) detection circuitry configured todetect a VSWR mismatch condition associated with the output loadterminal. The VSWR detection circuitry is usable by the controlcircuitry to set an isolation impedance of the quadrature coupler to atermination impedance range relative to the nominal output loadtermination impedance to improve isolation between the carrier and peakamplifiers for improving load insensitive load modulation operation.

Those skilled in the art will appreciate the scope of the disclosure andrealize additional aspects thereof after reading the following detaileddescription in association with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings incorporated in and forming a part of thisspecification illustrate several aspects of the disclosure, and togetherwith the description serve to explain the principles of the disclosure.

FIG. 1 is a schematic diagram of a conventional Doherty amplifier thatis related art.

FIG. 2 is a schematic diagram of a related art broadband reconfigurableDoherty power amplifier with tunable coupler.

FIGS. 3A and 3B are schematic diagrams of a related art multi-bandDoherty power amplifier.

FIG. 4 is a schematic of a related art inverted Doherty amplifier thatexchanges phases between a carrier amplifier and a peak amplifier.

FIG. 5A is a schematic diagram of a related art reconfigurable tunedDoherty amplifier.

FIG. 5B is a graph depicting ideal carrier amplifier transistorimpedance and peak amplifier current profiles for variouspeak-to-average power ratio (PAPR) type Doherty amplifiers.

FIG. 6A is a graph of drain efficiency versus normalized frequency for aconventional Doherty amplifier of the related art.

FIG. 6B is a graph of drain efficiency versus output power back off(OPBO) for a conventional Doherty amplifier of the related art.

FIG. 7 is a schematic diagram of a first embodiment that in accordancewith the present disclosure is a reconfigurable wideband load modulatedpower amplifier that is closed loop tunable.

FIG. 8 is a schematic diagram of a second embodiment that in accordancewith the present disclosure is a reconfigurable wideband load modulatedpower amplifier that is open loop tunable.

FIG. 9 is a schematic diagram of a third embodiment that in accordancewith the present disclosure is a reconfigurable wideband inverted phaseload modulated power amplifier that is closed loop tunable.

FIG. 10 is a schematic diagram of a fourth embodiment that in accordancewith the present disclosure is a reconfigurable wideband inverted phaseload modulated power amplifier that is open loop tunable.

FIG. 11 is a schematic diagram of a fifth embodiment that in accordancewith the present disclosure is a reconfigurable wideband inverted phaseload modulated power amplifier that is open loop tunable.

FIG. 12 is a graph depicting 4-port quadrature coupler impedancemodulation in accordance to the present disclosure.

FIG. 13 is a graph depicting 4-port quadrature coupler scatteringparameter S12 isolation between the carrier amplifier and the peakamplifier during operation with output impedance of 50Ω and a voltagestanding wave ratio (VSWR) of 1:1.

FIG. 14 is a graph depicting 4-port quadrature coupler scatteringparameter S12 isolation between the carrier amplifier and the peakamplifier during operation with an output impedance of 5Ω and a VSWR of10:1.

FIG. 15 is a graph depicting 4-port quadrature coupler scatteringparameter S12 isolation between the carrier amplifier and the peakamplifier during operation with an output impedance of 500Ω and a VSWRof 10:1.

FIG. 16 is a graph depicting gain versus power for a Lange-coupledmodulated amplifier during operation at isolation termination impedancesof 50Ω, 100Ω, 500Ω, 1000Ω, and 5000Ω.

FIG. 17 is a graph depicting gain versus power for a Lange-coupled loadmodulated amplifier during operation at isolation termination impedancesof 0Ω, 5Ω, 10Ω, 20Ω, 30Ω, 40Ω and 50Ω.

FIG. 18 is a graph depicting gain versus power for a Lange-coupledinverted load modulated amplifier during operation at isolationtermination impedances of 50Ω, 100Ω, 500Ω, 1000Ω, and 5000Ω.

FIG. 19 is a graph depicting gain versus power for a Lange-coupledinverted load modulated amplifier during operation at isolationtermination impedances of 0Ω, 5Ω, 10Ω, 20Ω, 30Ω, 40Ω and 50Ω.

FIG. 20 is a graph depicting gain versus power for a quadraturecoupled-line load modulated amplifier during operation at isolationtermination impedances of 50Ω, 100Ω, 500Ω, 1000Ω, and 5000Ω.

FIG. 21 is a graph depicting gain versus power for a quadraturecoupled-line load modulated amplifier during operation at isolationtermination impedances of 0Ω, 5Ω, 10Ω, 20Ω, 30Ω, 40Ω and 50Ω.

FIG. 22 is a graph depicting gain versus power for a quadraturecoupled-line inverted load modulated amplifier during operation atisolation termination impedances of 50Ω, 100Ω, 500Ω, 1000Ω, and 5000Ω.

FIG. 23 is a graph depicting gain versus power for a quadraturecoupled-line inverted load modulated amplifier during operation atisolation termination impedances of 0Ω, 5Ω, 10Ω, 20Ω, 30Ω, 40Ω and 50Ω.

FIG. 24 is a graph depicting gain versus power for a Lange-coupledinverted load modulated amplifier during operation at output frequenciesof 1.8 GHz, 2.15 GHz, and 2.5 GHz, ±15% bandwidth (BW).

FIG. 25 is a graph depicting gain versus power for a conventionalDoherty amplifier of the related art during operation at outputfrequencies of 1.8 GHz, 2.15 GHz, and 2.5 GHz, ±15% BW.

FIG. 26 is a graph depicting wideband output power and power addedefficiency (PAE) of an inverted load modulation embodiment of thepresent disclosure versus a conventional Doherty amplifier with bothoperating at 6 dB output power back off (OPBO).

FIG. 27 is a schematic diagram of a quadrature combined load modulatedpower amplifier.

FIG. 28 is a schematic diagram of an inverted phase quadrature combinedload modulated power amplifier.

FIG. 29 is a graph depicting measured quadrature phase load modulatedgain and efficiency versus output power.

FIG. 30 is a graph depicting measured quadrature inverted phase loadmodulated gain and efficiency versus output power.

FIG. 31 is a graph depicting conventional 50Ω quadrature operation forclass A, class B, and class C amplifiers with no load modulation.

FIG. 32 is an IV curve graph with loadlines depicting conventional classA (Rload) load line operation, low impedance (RLoad/2) load modulationunder lower power backed off signal, and low impedance (Rload/2) loadmodulation under low power backed off signal and lower supply voltageVq_m, illustrating more efficient load line impedance at lower supplyvoltage with higher current swing (than 2*Rload conventional DohertyOPBO load-line configuration).

FIG. 33 is a graph of load modulation combined with a Doherty amplifierconfigured for envelope tracking (ET).

FIG. 34 is a graph of measured total gain and collector efficiencyversus output power for a quadrature phase load modulated amplifier ofthe present disclosure illustrating improved backed off efficiencythrough the combination of load modulation (to lower Zc impedance ˜RL/2)and lower supply modulated operating voltage.

DETAILED DESCRIPTION

The embodiments set forth below represent the necessary information toenable those skilled in the art to practice the disclosure andillustrate the best mode of practicing the disclosure. Upon reading thefollowing description in light of the accompanying drawings, thoseskilled in the art will understand the concepts of the disclosure andwill recognize applications of these concepts not particularly addressedherein. It should be understood that these concepts and applicationsfall within the scope of the disclosure and the accompanying claims.

In this disclosure, a new type of load modulated amplifier is described.This new type of load modulated amplifier operates with a largedeparture from fundamental Doherty amplifier operation in three majoraspects. In particular, the phase of a carrier amplifier and peakamplifier making up the disclosed load modulated amplifier have beeninverted, while at the same time, the impedance modulation versus powermay be inverted for the carrier and peak amplifiers resulting in betterefficiency and peak to average power ratio (PAPR) while providingimproved bandwidth (BW) performance over conventional Dohertyamplifiers, and the modulated load impedance seen by either carrier orpeak amplifier can achieve a range that is as low as half of the nominalnormalized load impedance of the quadrature coupler, and as high astwice the nominal normalized load impedance of the quadrature coupler.It should be appreciated that conventional Doherty operationreconfigures the carrier amplifier load impedance to be as high as twicethe nominal normalized load impedance of the quadrature amplifier, butdoes not reconfigure the carrier amplifier below the nominal normalizedload impedance.

Because of the departure from Doherty operation of the impedance versuspower profile, where the carrier impedance is now decreasing withreduced input power level, and can produce a modulation load to eitherthe peak or carrier amplifier that is lower than the nominal normalizedload impedance of the quadrature coupler, and as low as half of thenominal normalized load impedance of the quadrature amplifier, whilemaintaining good output power back off (OPBO) efficiency, the amplifierembodiments of the present disclosure are not a type of Dohertyamplifier. In effect, the embodiments of the presently disclosedamplifier are novel and merit a new classification—an amplifier suigeneris.

In addition, the embodiments of the present disclosure enable a methodof preserving port-to-port (i.e., carrier to peak) amplifier isolationover a wide frequency bandwidth, which involves detecting a voltagestanding wave ratio (VSWR) and phase at the output, and then adjustingan isolation impedance on the coupler isolation port to provide improvedload insensitive and high efficiency operation, which is a fundamentalchallenge in Doherty amplifiers. Further still, this disclosure alsofocuses on a reduction to practice of the novel amplifier embodimentsdisclosed herein. The disclosed reduction to practice is a largedeparture from fundamental Doherty amplifier load operation in at leasttwo major aspects. The impedance modulation versus power is the oppositeof what is characteristic for a Doherty amplifier in which a carrieramplifier is introduced to relatively lower impedance. Another aspect isthat the impedance modulation that is presented to one or the otheramplifiers is lower than the nominal normalized impedance of thequadrature coupler. In particular, embodiments of the present disclosuresee half of the load impedance whereas a Doherty amplifier experiencestwice the load impedance during OPBO operation. As a result, an improvedsmall signal efficiency and PAPR is achieved while providing BWimprovements over traditional Doherty amplifiers. Moreover, since thepresent embodiments see half the load impedance, improvements inapplications having drain-voltage modulation techniques such as envelopetracking (ET), and average power tracking (APT) can be realized. Thepresent embodiments can also be used in place of traditional Dohertyamplifiers for enhanced operation of applications that includetraditional Doherty amplifiers.

FIG. 1 is a schematic of a related art Doherty amplifier 10 that isconfigured conventionally in a 2-way symmetrical design. One differencebetween the related art Doherty amplifier 10 and embodiments of thepresent disclosure is that a carrier amplifier 12 of the Dohertyamplifier 10 is communicatively coupled to an output terminal 14 througha 90° phase shifter 16. A peak amplifier 18 is coupled to an inputterminal 20 through another 90° phase shifter 22. A terminationimpedance 24 is coupled between the peak amplifier 18 and the outputterminal 14. Another difference is that the impedance presented to acarrier amplifier at ˜6 dB OPBO is twice the load impedance (i.e.,2*Rload). In the present disclosure, at least one of the embodiments hasa carrier amplifier that is coupled to the output through a zero degreecoupler. Moreover, at least one other embodiment of the presentdisclosure has an impedance presented to the carrier amplifier that isapproximately Rload/2 at roughly 6 dB OPBO or greater. This lowerimpedance results in higher carrier amplifier power efficiency when theamplifier is backed off from saturation. Note, that the load impedanceRload (not shown) is provided by a load that is usually coupled to theoutput terminal 14.

In particular, the embodiments of the present disclosure depart fromconventional Doherty amplifiers in at least the following ways. In atleast some of the disclosed embodiments, the carrier amplifier isdirectly coupled to the zero phase terminal through the ports of a4-port coupler, as opposed to coupling through a 90° phase shift to asumming node. Other embodiments are configured such that a terminationimpedance of the coupler is adjustable. Due to the termination impedanceadjustability, the carrier amplifier load impedance decreases as poweris backed off instead of increasing as occurs in a conventional Dohertyamplifier operation. Moreover, at least some of the disclosedembodiments are configured to detect a VSWR and automatically adjust theisolation termination impedance (Z_(ISO)) of the coupler to providemaximum power amplifier efficiency with good antenna VSWR load immunity.Further still, at least some embodiments are reconfigurable for loadmodulated amplification using an open loop signal that adjuststermination impedance for maximizing characteristics such as power,efficiency, linearity, bandwidth, load insensitivity or a combinationthereof.

FIG. 2 is a schematic of a related art reconfigurable broadband Dohertyamplifier 26 employing a 4-port quadrature coupler combiner 28 at anoutput load terminal 30 where load modulation of the carrier amplifieris achieved by changing a termination impedance 32 of a termination port34 to create Doherty amplifier operation. The termination impedance 32may be optimized for a given Pout, frequency or temperature, and mayinvolve continuous or discrete control from a control signal generatedby a detector 36. Note the coupled port polarity and impedance changewith power is inverted for the embodiments of the present disclosure. Asecond coupler 38 is coupled between an input terminal 40 and a peakamplifier 42 and a carrier amplifier 44. A second termination impedance46 is coupled to a second termination port 48.

FIGS. 3A and 3B are schematics of a related art low band amplifier 50and a high band amplifier 52, respectively, that each illustrates thebroadband nature of the use of Lange couplers 54, 56, 58, and 60. Morespecifically, the related art amplifiers 50 and 52 are configured foroperation at an operational frequency fo and 3fo and rely on theperiodic nature of the Lange quadrature couplers 54, 56, 58, and 60.Phasing during operation is such that the Lange couplers 54 and 56 eachprovide 90° quadrature characteristics at 1^(st) and roughly 3^(rd)harmonics with low loss. Thus, reconfigurability can be achieved inmultiple bands. Note that a direct coupled output 62 at 0° is coupled tothe peak amplifier 42 as in a conventional Doherty amplifier whichdeparts from embodiments of the present disclosure which are not limitedto Doherty amplifier operation. Also, a carrier impedance Zcarrier and apeak impedance Zpeak are >Zload for a load (not show) that is typicallycoupled to the output load terminal 30, whereas embodiments of thisdisclosure may utilize Zcarrier and Zpeak<Zload for achieving bothefficiency and load insensitivity.

FIG. 4 is a schematic of a related art inverted Doherty power amplifier64. The inverted Doherty power amplifier 64 saves area and thus cost formobile applications by eliminating matching circuitry. Basically, theinverted Doherty power amplifier 64 exchanges the phases between acarrier amplifier 66 and a peak amplifier 68. Note that a widerbandwidth can be obtained by eliminating output impedance transformer70, which in this case is a narrow-band impedance λ/4 transformer. Thisinherent feature is also preserved in embodiments of the presentdisclosure, since a 4-port quadrature coupler produces a wider andbroader band impedance transformation from the 50Ω output impedance. Asa result, there is no need for an output impedance transformer such asoutput impedance transformer 70 for the present embodiments, regardlessof the output phase configuration. The inverted Doherty power amplifier64 also includes a power splitter 72 coupled between an input IN and thecarrier amplifier 66 and the peak amplifier 68. Also, a narrow-band λ/4impedance inverter 74 that is communicatively coupled between outputs ofthe carrier amplifier 66 and the peak amplifier 68 through couplingcapacitors C1 and C2.

FIG. 5A is a schematic of a related art reconfigurable Doherty amplifier76 that in effect is tunable. The objective of this type of Dohertyamplifier is to increase the efficiency of the power amplifier over alarger PAPR by augmenting the load modulation of the reconfigurableDoherty amplifier 76 using matching networks 78 and 80 made up ofmicro-electro-mechanical systems (MEMs) tuning networks at the outputsof a carrier amplifier 82 and a peak amplifier 84. However, it should benoted that this is not a continuous operation. Although the range ofimpedances presented to the carrier amplifier 82 is much larger than aconventional Doherty amplifier, the reconfigurable Doherty amplifier 76still utilizes an increase of carrier load impedance as output power isbacked off as opposed to a reduction of carrier load impedance taughtfor operation of the embodiments of the present disclosure. Furthermore,as shown in FIG. 5B, the impedance modulation range practiced for thereconfigurable Doherty amplifier 76 shows impedances larger than R_(L)(50Ω), whereas the present embodiments may be configured to operate atimpedance modulations below and above R_(L). The reconfigurable Dohertyamplifier 76 also includes a power divider 86 coupled between a powerinput P_(IN) and the carrier amplifier 82 and the peak amplifier 84. Acombining network 88 is coupled between the matching networks 78 and 80,and the load R_(L).

FIGS. 6A and 6B are graphs for an enhanced related art Doherty amplifierusing drain modulation. In these examples, the supply or drain voltageof the carrier amplifier is under output power backed-off conditions inorder to further improve the output power efficiency. As shown, a largerOPBO efficiency range is one frequency bandwidth of operation that isalso improved by eliminating the output impedance transformer. However,a traditional quarter wave impedance inverter remains included and isdesigned to have broadband at only the peak efficiency backed-off powerlevel. A region shown inside a dashed box of FIG. 6B illustrates theimprovement in OPBO when carrier V_(DS1) supply is reduced compared topeak V_(DS2).

FIGS. 7-11 are schematics of embodiments of the present disclosure.Embodiments of the present disclosure differ from the related art in atleast the following significant ways. For one, the carrier amplifier ofthis disclosure in FIGS. 7-9 may be directly coupled to the zero phasetransmission ports of a 4-port coupler, as opposed to coupling through a90° phase shift to a summing node.

Moreover, the termination port impedance Z_(ISO) of the coupler isadjustable such that the carrier amplifier load impedance decreases aspower is backed off instead of increasing as in related art and therelated art discussed above. Further still, the present embodiments areconfigured to adjust the termination port impedance Z_(ISO) over a widerrange such that the carrier amplifier load impedance port of the couplermay see a range from Zload/2 to greater than 2*Zload, where Zload is thenominal output port impedance of the coupler. This extends thereconfigurable OPBO efficiency region of the load modulated amplifier.Even further, the embodiments of the present disclosure provide enhancedload insensitivity by detecting the output VSWR and adjusting thecoupler termination impedance above the nominal Zload impedance when thedetected VSWR mismatch is due to a lower than Zload mismatch impedance,and vice versa (for a non-inverting load modulation configuration). Thisflexibility improves port to port isolation between the carrieramplifier and the peak amplifier, which is a drawback of conventionalDoherty operation. Moreover, the Z_(ISO) tuning range mode may beselected from at least the choices of 0-Zload, or Zload-2*Zload forimproved load insensitive performance. Even further, load modulatedamplification is reconfigurable using an open loop signal that adjuststermination impedance for maximizing power, efficiency, linearity,bandwidth, load insensitivity or a combination thereof.

FIG. 7 is a schematic diagram of a first embodiment, that in accordancewith the present disclosure, is a reconfigurable load modulated poweramplifier 90 that is wideband and closed loop tunable. A carrieramplifier 92 is biased for class A operation and a peak amplifier 94 isbiased for class C operation. The carrier amplifier 92 and the peakamplifier 94 are coupled in parallel using a first quadrature coupler 96at an input 98 and a second quadrature coupler 100 at an output loadterminal 102. It should be noted that the bias of the peak amplifier 94may be set for other classes such as A or AB in order to configure theamplifier for a desired response. An input impedance tuning network 104is coupled between the first quadrature coupler 96 and ground. An outputimpedance tuning network 106 is coupled between the second quadraturecoupler 100 and ground. A detector 108 is a controller that generatessignals that tune the impedances of the input impedance tuning network104 and the output impedance tuning network 106 under continuouscontrol.

FIG. 8 is a schematic diagram of a second embodiment that in accordancewith the present disclosure is a reconfigurable wideband load modulatedpower amplifier 110 that is open loop tunable. In particular, thereconfigurable wideband load modulated power amplifier 110 includes anopen loop baseband signal look up table (LUT) 112 that is a controllerthat outputs baseband signals that include a first control signal S1 anda second control signal S2 that each are responsive to power, frequencyf0, and temperature T or any combination thereof.

In the present case the desired goal is to achieve high efficiency atOPBO by use of amplifier load modulation. The quadrature couplers may beimplemented as a Lange, a branch-line, or a coupled-line couplerconstruction or any other structure that provides a 90° phase split. Inthe embodiments shown in FIGS. 7 and 8, the polarity of the inputcouplers is such that the carrier amplifier 92 receives a signal that isin-phase with the input signal. The peak amplifier 94 receives a signalthat has a 90° phase shift with respect to the input signal. To maintainproper power combining at the output, the carrier amplified outputsignal undergoes a 90° phase shift before combining at a common outputnode (Z3), while the peak amplified signal is coupled to the output withzero or no phase shift. A termination port (Z4) is presented with avariable isolation termination impedance, Z_(ISO), which may be adjustedcontinuously or discretely to modulate a carrier amplifier loadimpedance (Z2) seen by the carrier amplifier 92 in order to improveefficiency as the output power is backed off of compression. Thisvariable tuning impedance means may be realized by asilicon-on-insulator (SOI) linear varistor for continuous or closed loopoperation. Alternatively, the variable tuning impedance may also berealized by a switched impedance network of inductors, resistors, andcapacitors using MEMs or SOI switch technologies. The input impedancetuning network 104 at a 4^(th) (isolation) port of the input quadrature4-port coupler may be tuned in conjunction with the output impedancetuning network 106 in order to adjust the amplitude and phase combiningof the carrier and peak amplifiers 92, 94 for best power performance.The impedance tuning networks 104 and 106 are controlled by the basebandsignals that are generated by the detector 108 (FIG. 7), or by basebandgenerated control signals S1 and S2 generated by the open loop basebandsignal LUT 112 (FIG. 8) that may be in response to output detectedpower, input detector power, DC current characteristics of the peakamplifier and the carrier amplifier, or output VSWR mismatchinformation.

Unlike a Doherty amplifier or the other related art discussed above, theembodiments of FIGS. 7 and 8 operate where Z_(ISO) is tuned such that Z2presented to the carrier amplifier 92 may be lower than the nominal loadimpedance, Zload (which is typically 50Ω), in order to obtain high OPBOefficiency. This is counter to the 2*Zload, in this case ˜100Ω,typically presented to the carrier amplifier at 6 dB OPBO for Dohertyoperation. Secondly, the embodiments of FIG. 7 and FIG. 8 practice arange of Z2 impedances (˜Zload/2 to ˜2*Zload that are presented to thecarrier amplifier 92 that is twice that of a conventional Dohertyamplifier. Thirdly, it is demonstrated in this disclosure that theisolation characteristics of the coupler between the carrier amplifierZ2 and peak amplifier Z1 may be improved under output VSWR mismatch byadjusting the Z_(ISO) impedance to be >Zload (50Ω) when the VSWRmismatch is due to an output impedance <Zload (50Ω) or vice versa.

A significant application is realized by a wide range of carrier andpeak amplifier load impedances that is used as a general load modulationmeans for improving power efficiency. For example, in conjunction withenvelope tracking (ET) or supply modulation techniques, where the supplyis reduced at lower power levels for the carrier amplifier to improvepower efficiency at OPBO, Z_(ISO) may be tuned such that the carrieramplifier load impedance Z2 is lower than Zload (50Ω) to provide anefficient load at the lower supply condition. The Z2 carrier load tuningprovides an extra degree of freedom to appropriately set the power andefficiency load line on the I-V curves of a power transistor of thecarrier amplifier 92. Moreover, the same technique may be applied to apower amplifier system where drain voltage and bias current are boththrottled down at lower powers to improve efficiency. The ability toadjust the load line can enable linear operation in addition toefficiency operation. This capability is very useful for improving thepower efficiency of gallium nitride-based devices and PAs whose high(˜5V) I-V knee voltage limits the efficiency performance when operatingat lower supply and drain voltages. The extra degree of freedom tomodulate the load impedance of the amplifier enables improvement inefficiency when the minimum practical drain voltage is reached in anenvelope tracking system.

FIGS. 9-11 show the 3^(rd), 4^(th) and 5th embodiments of the presentdisclosure. These embodiments are fundamentally different from the1^(St) and 2^(nd) embodiments in the phase combining of the carrieramplifier 92 and the peak amplifier 94 and depart from the phasecombining of the Doherty amplifier operation in general. Beginning withFIG. 9, a reconfigurable wideband inverted phase load modulated poweramplifier 114 that is closed loop tunable is shown. In particular, thecarrier amplifier 92 biased in class A and the peak amplifier 94 biasedin class C are in a combined parallel connection using a firstquadrature coupler 116 at the input and a second quadrature coupler 118at the output load terminal 102. It should be noted that the bias of thepeak amplifier 94 may be set for other classes such as A or AB in orderto configure the amplifier for a desired response. In the present casethe desired goal is to achieve high efficiency at OPBO by use ofamplifier load modulation. The quadrature couplers 116 and 118 may beimplemented as a Lange, a branch-line, or a coupled-line couplerconstruction or any other structure that provides a 90 degree phasesplit. In these 3^(rd), 4^(th) and 5^(th) embodiments, the polarity ofthe input couplers are different than in the 1^(St) and 2^(nd)embodiments, and are such that the carrier amplifier 92 receives asignal that is phase shifted 90 degrees with respect to the inputsignal. The peak amplifier 94 receives a signal that is in-phase (0degree phase shifted) with respect to the input signal.

To maintain proper power combining at the output, the carrier amplifiedoutput signal undergoes zero or no phase shift before combining at thecommon output node (Z3), while the peak amplified signal is coupled tothe output with 90 degree phase shift. Unlike the 1^(st) and 2^(nd)embodiments and the related art discussed above, the termination port(Z4) is presented with a variable isolation termination impedance,Z_(ISO), which may be adjusted continuously or discretely to modulatethe carrier amplifier load impedance (Z1) seen by the carrier amplifier92 in order to improve efficiency as the output power is backed off ofcompression. This variable tuning impedance means may be realized by anSOI linear varistor for continuous or closed loop operation. Thevariable tuning impedance may also be realized by a switched impedancenetwork of inductors, resistors, and capacitors usingmicro-mechanical-systems (MEMs) switch networks or SOI switchtechnologies. The input impedance tuning network 104 at the 4^(th)(isolation) port of the input quadrature 4-port coupler may be tuned inconjunction with the output impedance tuning network 106 in order toadjust the amplitude and phase combining of the carrier and peakamplifiers 92, 94 for best power performance. The impedance tuningnetworks 104, 106 may be controlled with a control signal that isgenerated by the detector 108, a look up table (LUT) or basebandgenerated signals (4^(th) embodiment) that may be in response to outputdetected power, input detected power, DC current characteristics of thepeak amplifier 94 and the carrier amplifier 92, or output VSWR mismatchinformation (5^(th) embodiment). For example, FIG. 10 is a schematicdiagram of the 4^(th) embodiment that in accordance with the presentdisclosure is a reconfigurable wideband inverted phase load modulatedpower amplifier 120 that is open loop tunable by way of the open loopbaseband signal LUT 112.

Unlike a Doherty amplifier, the related art discussed above, or the1^(st) and 2^(nd) embodiments, the 3^(rd), 4^(th) and 5^(th) embodimentsoperate wherein firstly Z_(ISO) is tuned such that Z1 presented to thecarrier amplifier 92 may be lower than the nominal load impedance, Zload(which is typically 50Ω), in order to obtain high OPBO efficiencycharacteristics. Presenting Z1 lower to the carrier amplifier 92 iscounter to the ˜2*Zload (˜100Ω) typically presented to the carrieramplifier at 6 dB OPBO for Doherty operation. Secondly, the 3^(rd),4^(th) and 5^(th) embodiments are practicing a range of Z1 impedances(˜Zload/2 to ˜2*Zload, FIG. 12) presented to the carrier amplifier orpeak amplifier that is twice that of a conventional Doherty amplifier.Thirdly, it is demonstrated in this disclosure that the isolationcharacteristics of the coupler between the carrier amplifier Z1 and peakamplifier Z2 may be improved under output VSWR mismatch by adjusting theZ_(ISO) impedance to be >Zload (50Ω) when the VSWR mismatch is due to anoutput impedance <Zload (50Ω) or vice versa.

A significant application of the embodiments of the present disclosureis that a wide range of the carrier amplifier and the peak amplifierload impedances may be used as a general load modulation means forimproving power efficiency. For example, in conjunction with envelopetracking or supply modulation techniques where the supply is reduced atlower power levels for the carrier amplifier to improve power efficiencyat OPBO, Z_(ISO) may be tuned such that the carrier amplifier loadimpedance Z1 is lower than Zload (50Ω) to provide an optimized load atthe lower supply condition. This Z1 carrier load tuning provides anextra degree of freedom to tune the power and efficiency load line onthe I-V curve of the power transistor of the carrier amplifier 92 for adesired performance. Moreover, the same technique may be applied to apower amplifier system where drain voltage and bias current are boththrottled down at lower powers to improve efficiency. The ability toadjust the load line can enable linear operation in addition toefficiency operation. This capability is very useful for improving thepower efficiency of gallium nitride-based devices and power amplifierswhose high (˜5V) I-V knee voltage limits the efficiency performance whenoperating at lower supply and drain voltages. The extra degree offreedom to modulate the load impedance of the amplifier enablesimprovement in efficiency when the minimum practical drain voltage isreached in an envelope tracking system.

FIG. 11 is a schematic diagram of a 5^(th) embodiment that in accordancewith the present disclosure is a reconfigurable wideband inverted phaseload modulated power amplifier 124 that is open loop tunable. This5^(th) embodiment shows an application of the 3^(rd) and 4^(th)embodiments, but is also applicable to the 1^(St) and 2^(nd) embodimentsfor improving the load-insensitivity of the load modulated poweramplifier under high VSWR conditions such as when there is a mismatchwith an antenna coupled to the output of the load modulated poweramplifier. This 5^(th) embodiment shows a system where the output VSWRis detected through VSWR detector 126 that includes circuitry thatdetermines the mismatched load impedance. For example, the VSWRdetection circuitry can detect real and imaginary or S-parametermagnitude and angle or other VSWR related parameters. This informationis usable to determine whether the VSWR is due to a mismatched loadimpedance (Zload) that is >50Ω (nominal design) or <50Ω in its simplestpractice.

The characteristic nature of a quadrature coupler which is illustratedin FIGS. 13, 14, and 15, indicate that when the output is perfectlymatched to 50Ω and Z_(ISO) is 50Ω, the isolation between the couplerimpedance ports Z1 and Z2, corresponding to the isolation between thecarrier amplifier 92 and the peak amplifier 94, is excellent, asillustrated in FIG. 13. When the output port (Z3) is mismatched with a5Ω output load corresponding to a 10:1 VSWR mismatched condition, thenthe isolation degrades when Z_(ISO) is 50Ω or less. However, if Z_(ISO)is increased to a value >50Ω, the isolation between ports Z1 and Z2improves dramatically. When the output port (Z3) is mismatched with a500Ω output load corresponding to a 10:1 VSWR mismatched condition, thenthe isolation degrades when Z_(ISO) is 50Ω or greater. However, ifZ_(ISO) is decreased to a value <50Ω, the isolation between ports Z1 andZ2 improves dramatically. It is illustrated in FIGS. 16-23 that improvedOPBO efficiency may be achieved in either region where Z_(ISO) is >50Ωor Z_(ISO)<50Ω. Therefore, a means for providing load insensitivity andhigh OPBO efficiency may be achieved by detecting the VSWR mismatchcondition and tuning Z_(ISO) in the appropriate region for maximumefficiency:

-   -   1) When VSWR detector measures Zload>50Ω, Z_(ISO) is optimized        for <50Ω    -   2) When VSWR detector measures Zload<50Ω, Z_(ISO) is optimized        for >50Ω        This may be implemented through a pre-calibrated LUT that        contains optimum efficiency solutions for Z_(ISO) as a function        of (Pin, frequency, VSWR mismatch information) as an example.

An exemplary embodiment of the present disclosure was modeled andsimulated in a gallium nitride (GaN) high electron mobility transistor(HEMT) technology with a transition frequency fT˜90 GHz and off-statebreakdown voltage of ˜50V. The knee voltage of the I-V characteristicsof the technology is ˜5V while the nominal supply voltage is no greaterthan 12V at an approximate current density of 500 mA/mm of gateperiphery. Embodiments of this disclosure were designed using AWRCorporation's microwave office (MWO) simulation tools with a foundryprocess design kit (PDK).

FIG. 12 shows the 4-port quadrature coupler impedance modulation of thecarrier amplifier 92 (Z2, zero degree) and peak amplifier 94 (Z1, 90degree) amplifier load impedance ports as a function of the isolationtermination impedance Z_(ISO). It should be recognized that Z_(ISO) atZ4 may also be complex impedance. For simplicity Z2 and Z1 loadimpedance characteristics are shown for swept real values of Z_(ISO) forvarious quadrature couplers, 1) Surface Mount Branch-line, 2) Coupledline, and 3) Lange. The Lange coupler has the widest frequency bandwidthcapability and also shows the widest range of the carrier amplifier (Z2)and the peak amplifier (Z1) port impedance tuning range. The impedancetuning range for Z2 may vary from Zload/2 to greater than 2*Zload whereZload is the normalized 50Ω coupler characteristic impedance. This rangeis 2 times larger than a conventional symmetric 2-way Doherty impedanceinverter which typically ranges from Zload up to 2*Zload at 6 dB backoff power. Therefore, the embodiments of the present disclosure provideboth two times greater load modulation as well as the ability tomodulate the impedance below Zload, unlike the conventional Dohertyimpedance inverter operation. Moreover, the present disclosuredemonstrates that high efficiency may be achieved by inverting thedirection of the impedance modulation where the carrier amplifier loadZ2 is tuned for lower impedance than Zload at output power back off(OPBO) (Z_(ISO)<Zload) while the peak amplifier 94 is tuned for highimpedance Z1 at OPBO which is the opposite of the Doherty operation.Therefore, embodiments of this disclosure can be assigned to a moregeneral load modulated amplifier class.

FIG. 13 is a graph illustrating that when the output is matched, thebest carrier-peak amplifier isolation occurs when Z_(ISO) is equal toZload=Zout=50Ω. At the extreme Z_(ISO) impedances of 0.1Ω and 1K Ω, theisolation is still reasonably better than 7 dB across a wide band but isnot ideal. However, it is still better than the isolation a quarter-waveimpedance inverter would be able to achieve across this 2 octavefrequency band or even a more modest 2-2.7 GHz cellular applicationband. Under extreme output VSWR mismatches it will be shown that Z_(ISO)may be chosen to improve the 2 octave broadband isolation by choosingone of two ranges of Z_(ISO): 1) Z_(ISO)>Zload or 2) Z_(ISO)<Zload(50Ω).

FIG. 14 is a graph showing that when the output has a VSWR of 10:1 dueto a Zout impedance of 5Ω (Zout<50Ω), a Z_(ISO) in the range of >50Ω maybe chosen for improved isolation and optimum efficiency performance.Later in the disclosure it will be shown that a Z_(ISO) may be chosen ineither Z_(ISO)>50Ω or Z_(ISO)<50Ω ranges to achieve the enhanced OPBOefficiency, counterintuitive to Doherty amplifier operation, andtherefore this disclosure is referred to more generally as a loadmodulation amplifier.

FIG. 15 is a graph illustrating that when the output has a VSWR of 10:1due to a Zout impedance of 500Ω (Zout>50Ω), a Z_(ISO) in the range of<50Ω may be chosen for improved isolation and optimum efficiencyperformance. Later in the disclosure it will be shown that a Z_(ISO) maybe chosen in either Z_(ISO)>50Ω or Z_(ISO)<50Ω to achieve the enhancedOPBO efficiency, counterintuitive to Doherty operation and conventionalwisdom.

FIG. 16 is a graph depicting gain and power added efficiency (PAE)versus Pout for the Lange-coupled load modulated amplifier of the 1^(St)and 2nd embodiments where the isolation impedance Z_(ISO) is swept from50Ω to 5KΩ. Also shown as a reference comparison is the gain and PAEversus Pout for a conventional 2-way Doherty Amplifier. As Z_(ISO) isincreased the Z2 load presented by the coupler to the carrier amplifieralso increases and results in a higher increase in PAE as the outputpower is backed off. Note that the gain is flat, exhibiting reasonablylow AM-AM distortion. The AM-PM distortion (not shown) is also wellbehaved and is less than a couple of degrees over the applicable outputpower range. Also note that these amplifiers were designed with linearoperation as a focus.

FIG. 17 is graph illustrating gain and PAE versus Pout for theLange-coupled load modulated amplifier of the 1^(St) and 2^(nd)embodiments (FIGS. 7 and 8), where the isolation impedance Z_(ISO) isswept from 50Ω to 0Ω. Also shown as a reference comparison is the gainand PAE versus Pout for a conventional 2-way Doherty amplifier. AsZ_(ISO) is decreased, the Z2 load presented by the coupler to thecarrier amplifier also decreases, and results in a higher increase inPAE as the output power is backed off. This mode of operation departsfrom the traditional Doherty operation where the carrier amplifier ispresented a lower impedance (˜Zload/2) at OPBO and actually resulted inslightly better OPBO efficiency than when the carrier amplifier waspresented a large impedance (˜2*Zload) under OPBO operation (Dohertytype amplifier). This is believed to be due to the complexcharacteristics of the quadrature coupler combined with the carrieramplifier and peak amplifier bias settings where a more favorablecomplex load line is provided. Note that the gain is still flat,exhibiting reasonably low AM-AM distortion. The AM-PM distortion (notshown) is also well behaved and is less than a couple of degrees overthe applicable output power range. Also note that these amplifiers weredesigned with linear operation as a focus.

FIG. 18 is a graph depicting gain and PAE versus Pout for theLange-coupled inverted load modulated amplifier of the 3^(rd) through5^(th) embodiments (FIGS. 9-11) where the isolation impedance Z_(ISO) isswept from 50Ω to 5KΩ. Also shown as a reference comparison is the gainand PAE versus Pout for a conventional 2-way Doherty amplifier. AsZ_(ISO) is increased, the Z1 load presented by the coupler 118 to thecarrier amplifier 92 decreases and results in a higher increase in PAEas the output power is backed off. This mode of operation departs fromthe traditional Doherty amplifier operation where the carrier amplifieris presented a lower impedance (˜Zload/2) at OPBO and actually resultedin slightly better OPBO efficiency than when the carrier amplifier 92was presented a large impedance (˜2*Zload) under OPBO operation (Dohertytype amplifier) as shown in FIG. 19. This is believed to be due to thecomplex characteristics of the quadrature coupler combined with thecarrier amplifier and peak amplifier bias settings where a more optimumcomplex load line is provided. Note that the gain is still flat,exhibiting reasonably low AM-AM distortion. The AM-PM distortion (notshown) is also well behaved and is less than a couple of degrees overthe applicable output power range. Also note that the amplifiers 114,120, and 124 were designed with linear operation as a focus.

FIG. 19 is a graph illustrating gain and PAE versus Pout for theLange-coupled inverted load modulated amplifier of the 3^(rd) through5^(th) embodiments (FIGS. 9-11), where the isolation impedance Z_(ISO)is swept from 0Ω to 50Ω. Also shown as a reference comparison is thegain and PAE versus Pout for a conventional 2-way Doherty amplifier. AsZ_(ISO) is decreased, the Z1 load presented by the coupler 118 to thecarrier amplifier 92 increases and results in a higher increase in PAEas the output power is backed off. Note that the gain is flat,exhibiting reasonably low AM-AM distortion. The AM-PM distortion (notshown) is also well behaved and is less than a couple of degrees overthe applicable output power range. Also note that the amplifiers 114,120, and 124 were designed with linear operation as a focus.

FIG. 20 is a graph showing gain and PAE versus Pout for the quadraturecoupled line load modulated amplifier of the 1^(St) and 2^(nd)embodiments (FIGS. 7 and 8) where the isolation impedance Z_(ISO) isswept from 50Ω to 5KΩ. Also shown as a reference comparison is the gainand PAE versus Pout for a conventional 2-way Doherty. As Z_(ISO) isincreased the Z2 load presented by the coupler to the carrier amplifieralso increases and results in a higher increase in PAE as the outputpower is backed off. Note that the gain is flat exhibiting reasonablylow AM-AM distortion. The AM-PM distortion (not shown) is also wellbehaved and is less than a couple of degrees over the applicable outputpower range. Also note that the amplifiers 90 and 110 were designed withlinear operation as a focus.

FIG. 21 is a graph illustrating gain and PAE versus Pout for thequadrature coupled line load modulated amplifier of the 1^(St) and2^(nd) embodiments (FIGS. 7 and 8) where the isolation impedance Z_(ISO)is swept from 50Ω to 0Ω. Also shown as a reference comparison is thegain and PAE versus Pout for a conventional 2-way Doherty amplifier. AsZ_(ISO) is decreased, the Z2 load presented by the coupler to thecarrier amplifier also decreases and results in a higher increase in PAEas the output power is backed off. This mode of operation departs fromthe traditional Doherty operation where the carrier amplifier ispresented a lower impedance (˜Zload/2) at OPBO and actually resulted inslightly better OPBO efficiency than when the carrier amplifier waspresented a large impedance (˜2*Zload) under OPBO operation (Dohertytype). This is believed to be due to the complex characteristics of thequadrature coupler combined with the carrier amplifier and peakamplifier bias settings where a more optimum complex load line isprovided. Note that the gain is still flat exhibiting reasonably lowAM-AM distortion. The AM-PM distortion (not shown) is also well behavedand is less than a couple of degrees over the applicable output powerrange. Also note that the amplifiers 90 and 110 were designed withlinear operation as a focus.

FIG. 22 illustrates gain and PAE versus Pout for the quadraturecoupled-line inverted load modulated amplifier of the 3^(rd) through5^(th) embodiments (FIGS. 9-11) where the isolation impedance Z_(ISO) isswept from 50Ω to 5KΩ. Also shown as a reference comparison is the gainand PAE versus Pout for a conventional 2-way Doherty amplifier. AsZ_(ISO) is increased, the Z1 load presented by the coupler to thecarrier amplifier decreases and results in a higher increase in PAE asthe output power is backed off. This mode of operation departs from thetraditional Doherty amplifier operation where the carrier amplifier ispresented a lower impedance (˜Zload/2) at OPBO and actually resulted inslightly better OPBO efficiency than when the carrier amplifier waspresented a large impedance (˜2*Zload) under OPBO operation (Dohertytype) as shown in the upcoming FIG. 29. This is believed to be due tothe complex characteristics of the quadrature coupler combined with thecarrier amplifier and peak amplifier bias settings where a more optimumcomplex load line is provided. Note that the gain is still flat,exhibiting reasonably low AM-AM distortion. The AM-PM distortion (notshown) is also well behaved and is less than a couple of degrees overthe applicable output power range. Also note that the amplifiers 114,120, and 124 were designed with linear operation as a focus.

FIG. 23 illustrates gain and PAE versus Pout for the quadrature coupledline inverted load modulated amplifier of the 3^(rd) through 5^(th)embodiments (FIGS. 9-11) where the isolation impedance Z_(ISO) is sweptfrom 0Ω to 50Ω. Also shown as a reference comparison is the gain and PAEversus Pout for a conventional 2-way Doherty amplifier. As Z_(ISO) isdecreased, the Z1 load presented by the coupler to the carrier amplifierincreases and results in a higher increase in PAE as the output power isbacked off. Note that the gain is flat, exhibiting reasonably low AM-PMdistortion. The AM-PM distortion (not shown) is also well behaved and isless than a couple of degrees over the applicable output power range.Also note that the amplifiers 114, 120, and 124 were designed withlinear operation as a focus.

FIG. 24 is a graph illustrating Lange-coupled inverted load modulatedamplifier efficiency and gain response at 1.8, 2.15, and 2.5 GHz. Thegraph traces are for the isolation impedance Z_(ISO)=5KΩ when thecarrier amplifier 92 sees a small load impedance ˜Zload/2. Over thefrequency range of +/−15% and a back off power range of 6 dB, theinverted load modulated amplifier achieves roughly better than 39%efficiency across the band. As expected, the efficiency is poorest atthe upper band frequency of 2.5 GHz. The upper band efficiency may beimproved by adding inductive impedance to Z_(ISO) in series with aresistive load impedance Rload.

FIG. 25 is a graph illustrating the Doherty amplifier efficiency andgain response at 1.8, 2.15, and 2.5 GHz. Over the frequency range of+/−15% and a back off power of 6 dB, the conventional Doherty amplifierachieves no better than 38% efficiency at the lower band of 1.8 GHz andonly 29% at the upper band frequency of 2.5 GHz. This is roughly a 10%degradation over the inverted load modulated power amplifiers of thisdisclosure over the +/−15% bandwidth.

FIG. 26 is a graph showing the wideband response of the inverted loadmodulated amplifier versus the conventional Doherty amplifier over a+/−15% bandwidth at an OPBO of ˜6 dB for each amplifier respectively. Inparticular, FIG. 26 illustrates the improved 6 dB OPBO PAE of theinverted load modulated amplifier embodiment which maintains >39% PAEover a wide band with 1 dB more 6-dB OPBO Pout while the conventionalDoherty amplifier has as much as 14% and on average >10% poorer 6-dBOPBO PAE over the frequency range.

FIG. 27 is a schematic diagram of a quadrature combined load modulatedpower amplifier 128. The terminating impedance of the quadrature coupler100 coupled to the output load terminal 102 may be dynamically changed(continuous or discrete steps) to present a desired load impedance toeither the carrier amplifier 92 or the peak amplifier 94 for maximizingpower, efficiency or linearity at a given power level, frequency, ortemperature. The input termination port impedance may also be tunedsimultaneously to obtain optimum RF performance. Note that the phaserelationship of the carrier amplifier 92 in relation to the output is 90degrees offset, which is similar to the Doherty amplifier. What isdifferent is that in this particular embodiment, the carrier amplifier92 will see a load impedance which is lower than Zload at some OPBOlevel below the overall amplifier saturation point, and this is achievedby tuning Z_(ISO)<Zload. This is contrary to the operating principles ofthe Doherty amplifier whose carrier load impedance increases from Rloadto 2*Rload at OPBO. Moreover, the carrier amplifier 92 may see animpedance that is Zload/2 at OPBO below the saturated power level of thecarrier amplifier 92. It should be noted that the Doherty amplifiercannot provide a carrier load impedance below Rload and therefore isrestricted in load modulation range. The result of low carrier loadimpedance at OPBO is illustrated in upcoming FIG. 29.

FIG. 28 is a schematic diagram of an inverted phase quadrature combinedload modulated power amplifier 130. This is different from the previousembodiment of FIG. 27, in that the output of the carrier amplifier 92 iscoupled in-phase at a combined amplifier output. The terminatingimpedance of the output coupler may be dynamically changed (continuousor discrete steps) to present a desired load impedance to either thecarrier amplifier 92 or the peak amplifier 94 for optimizing power,efficiency, or linearity at a given power level, frequency, ortemperature. The input termination port impedance may also be tunedsimultaneously to obtain optimum RF performance. Note that the phaserelationship of the carrier amplifier 92 in relation the output is now 0degrees offset, unlike the Doherty amplifier. Like the previousembodiment, the carrier amplifier 92 will see a load impedance which islower than Zload at some OPBO level below the overall amplifier'ssaturation point, and this is achieved by tuning Z_(ISO)>Zload. This iscontrary to the operating principles of the Doherty amplifier whosecarrier load impedance increases from Rload to 2*Rload at OPBO.Moreover, the carrier amplifier 92 may see an impedance that is Zload/2at OPBO below the saturated power level of the carrier amplifier 92. Itshould be noted that the Doherty amplifier cannot provide a carrier loadimpedance below Rload and therefore is restricted in load modulationrange. The result of low carrier load impedance at OPBO is illustratedin upcoming FIG. 30.

FIG. 29 is a graph of measured gain and collector efficiency vs. Poutfor the quadrature phase load modulated power amplifier 128 of FIG. 27.The carrier amplifier 92 is biased in class A and the peak amplifier isbiased in class C. These characteristics were measured for Z_(ISO)=50Ωand Z_(ISO)=0Ω to illustrate the range of load modulation between theseZ_(ISO) impedances, but should be noted that a continuous or discreterange of impedances may be employed. These impedances correspond to acarrier amplifier load impedance of 50Ω and 25Ω respectively. For thelower carrier impedance of 25Ω, a dramatic improvement in OPBOefficiency is obtained compared to the 50Ω carrier impedance setting.This operating principle is contrary to the Doherty amplifier, whichmodulates the carrier load to higher impedances at ˜6 dB OPBO toincrease the efficiency. The fundamental difference is that the Dohertyamplifier is setting the OPBO load for maximum saturated class Aoperation, whereas the present disclosure places the operation of thecarrier amplifier 92 in the linear region.

FIG. 30 is a graph of measured gain and collector efficiency versus Poutfor the inverted quadrature phase load modulated power amplifier 130 ofFIG. 28. The carrier amplifier 92 is biased in class A and the peakamplifier 94 is biased in class C. These characteristics were measuredfor Z_(ISO)=50Ω and Z_(ISO)=5KΩ but should be noted that a continuous ordiscrete range of impedances may be employed. These impedancescorrespond to a carrier amplifier load impedance of 50Ω and 25Ωrespectively. For the lower carrier impedance of 25Ω, a dramaticimprovement in OPBO efficiency is obtained compared to the 50Ω carrierimpedance setting. This operating principle is contrary to the Dohertyamplifier, which modulates the carrier load to higher impedances at ˜6dB OPBO to increase the efficiency. The fundamental difference is thatthe Doherty amplifier is setting the OPBO load for maximum saturatedclass A operation, whereas the present disclosure places the amplifierin the linear region.

As a reference comparison, FIG. 31 is a graph that shows the measuredconventional 50Ω quadrature operation of the quadrature combined loadmodulated power amplifier 128 where both the carrier amplifier 92 andthe peak amplifier 94 are operated at the same bias and Z_(ISO)=50Ω. Thecorresponding efficiency and gain curves for class A, B, and C areillustrated and show the expected characteristic efficiency ‘drop-off’response as output power is backed off. The plots show thecharacteristic gain expansion of the class C case and its associatedhigher efficiency than the class B or C cases as power ramps up. At ˜5-6dB OPBO, or an output power of ˜36 dBm, the efficiency is at best 30%for the class C case and 25% for the class A case, as compared to 38 and40% for the quadrature phase and inverted phase load modulatedamplifiers at the lower load impedance. The lower impedance capabilityof this new load modulation amplifier has far reaching implications tohigh linear efficiency optimization at lower voltages. It may becombined with envelope tracking or drain mode biasing such as class G inorder to optimize power efficiency further. And this would come withother inherent benefits such as wider bandwidth and real timereconfigurability.

FIG. 32 is a graph illustrating current-voltage I-V load line operationof the carrier amplifier 92 when the amplifier is backed off of itssaturation. In particular, the IV curve graph includes load-linesdepicting conventional class A (Rload) load line operation, lowimpedance (RLoad/2) load modulation under lower power backed off signal,and low impedance (Rload/2) load modulation under low power backed offsignal and lower supply voltage Vq_m, illustrating more efficient loadline impedance at lower supply voltage with higher current swing (than2*Rload). The voltage and current excursion for a conventional loadmodulation amplifier such as a Doherty amplifier is constricted by the2*Rload impedance and is limiting in power efficiency for smaller thansaturated power operation. In order to improve the linear efficiency ofthe carrier amplifier 92 at backed off powers below saturation for thecarrier amplifier, the load impedance may be reduced as we aresuggesting in the present disclosure. In the embodiments described inthis disclosure, higher linear power can be obtained using a loadimpedance of Rload/2 resulting in a root mean squared current Irmshaving a larger excursion for the same DC power. This results inimproved small signal power efficiency as compared to the larger load.In addition, drain voltage could be reduced via APT or ET (as examples)in conjunction with lowering the load impedance in order to obtain evenhigher small signal power efficiency illustrated in thick dashed line.One approach is to reduce the Rp load towards ˜Rload/2 to improve the RFIds swing for a given RF Vout swing. This will increase thePav=Vrms*Irms by a large factor due to the larger AC current swing(assuming that gain is large). The factor of four reduction in Rloadwill result in a factor of four increase in RF Ids for the given RF Voutswing. This will require a larger RF Vin swing, however, if there issufficient amplifier gain, then this should be negligible and linearefficiency will be improved dramatically. Since the RF Vout voltage isnot clipped by the knee, further improvement in drain efficiency may beachieved by reducing the quiescent Vdrain or voltage applied to theamplifier (via average power tracking (APT) or ET for example). Thepotential improvement is conceptually illustrated for increasing theefficiency of a related art Doherty amplifier designed for envelopetracking (ET).

FIG. 33 is a graph that illustrates how load modulation may improve theefficiency of a Doherty amplifier designed for ET. It is previouslyknown that ET can be used to improve the >6 dB backed off efficiency ofa conventional Doherty amplifier. At 6 dB OPBO, the peak amplifier isoff and the carrier amplifier is presented with a 2*Ropt class A loadand is saturated. As output power is backed off even further, thecarrier amplifier becomes uncompressed and its efficiency starts todecrease. ET or APT may be applied to recover the efficiency by loweringthe drain voltage of the carrier amplifier as conceptually illustratedby the efficiency envelope of FIG. 33 shown in thick black line. Tofurther improve the efficiency, the load of the carrier may be reducedin order to allow more Irms excursion and achieve better Pout for agiven drain voltage bias point depicted as a bold dotted line. Thepresent load modulation amplifier embodiments disclosed herein canenable this type of operation while providing reconfigurability overwide modulation bandwidths, power ranges, and carrier frequency ofoperation.

FIG. 34 is a graph of measured total gain and collector efficiencyversus output power for a quadrature phase load modulated amplifier ofthe present disclosure illustration improved backed off efficiencythrough the combination of load modulation (to lower Zc impedance ˜RL/2)and lower supply modulated operating voltage. Moreover, FIG. 34 depictspreliminary data indicating the benefits that can be achieved bycombining the load modulation amplifier embodiments of this disclosurewith drain-voltage modulation at low powers in order to further improvethe OPBO efficiency. It should be noted that this is a breadboarddemonstration using available amplifier cells, and that a printedcircuit board (PCB) used for bread-boarding and the breadboard tuningwas not designed for maximum efficiency or power. It is also believedthat the inverted phase load modulation amplifier, such as those shownin FIGS. 9-11, will achieve less amplitude modulation—amplitudemodulation (AM-AM) distortion than will the reconfigurable loadmodulated power amplifier 90 of FIG. 7. The measured data of FIG. 34 isvery encouraging in light of this and demonstrates the fundamentallinear efficiency improvement that can be obtained from combining lowimpedance load modulation with low drain voltage operation.

Those skilled in the art will recognize improvements and modificationsto the embodiments of the present disclosure. All such improvements andmodifications are considered within the scope of the concepts disclosedherein and the claims that follow.

What is claimed is:
 1. A reconfigurable load modulation amplifiercomprising: a carrier amplifier; a peak amplifier coupled in parallelwith the carrier amplifier; and a quadrature coupler configured tocombine power from both the carrier amplifier and the peak amplifier foroutput through an output load terminal.
 2. The reconfigurable loadmodulation amplifier of claim 1 wherein the carrier amplifier is coupledto the output load terminal through 0° phase shift ports of thequadrature coupler.
 3. The reconfigurable load modulation amplifier ofclaim 1 wherein the carrier amplifier is coupled to the output loadterminal through 90° phase shift ports of the quadrature coupler.
 4. Thereconfigurable load modulation amplifier of claim 1 further includingcontrol circuitry coupled to an isolation port of the quadrature couplerand configured to provide adjustable impedance at the isolation port ofthe quadrature coupler.
 5. The reconfigurable load modulation amplifierof claim 4 wherein the control circuitry comprises a switch network thatis configured to receive a control signal for adjusting an impedancepresented to the isolation port of the quadrature coupler.
 6. Thereconfigurable load modulation amplifier of claim 5 wherein the switchnetwork comprises a micro-electro-mechanical systems (MEMs) switchnetwork.
 7. The reconfigurable load modulation amplifier of claim 5wherein the switch network is a silicon-on-insulator (SOI) switchnetwork.
 8. The reconfigurable load modulation amplifier of claim 4wherein impedance at the isolation port of the quadrature coupler istunable to impedance values above a terminal load impedance at theoutput load terminal.
 9. The reconfigurable load modulation amplifier ofclaim 4 wherein impedance at the isolation port of the quadraturecoupler is tunable to impedance values below a terminal load impedanceat the output load terminal.
 10. The reconfigurable load modulationamplifier of claim 4 wherein impedance at the isolation port of thequadrature coupler is tunable such that the carrier amplifier ispresented with a quadrature coupler load impedance that is below aterminal load impedance at the output load terminal.
 11. Thereconfigurable load modulation amplifier of claim 4 wherein impedance atthe isolation port of the quadrature coupler is continuously tunablesuch that the carrier amplifier is presented with a quadrature couplerload impedance that ranges from around about half an output loadtermination impedance to around about twice the output load terminationimpedance.
 12. The reconfigurable load modulation amplifier of claim 4wherein impedance at the isolation port of the quadrature coupler iscontinuously tunable such that the peak amplifier is presented with aquadrature coupler load impedance that ranges from around about half anoutput load termination impedance to around about twice the output loadtermination impedance.
 13. The reconfigurable load modulation amplifierof claim 10 wherein the quadrature coupler load impedance is aroundabout half the terminal load impedance at the output load terminal. 14.The reconfigurable load modulation amplifier of claim 4 whereinimpedance at the isolation port of the quadrature coupler is tunablesuch that the carrier amplifier is presented with a quadrature couplerload impedance that is above a terminal load impedance at the outputload terminal.
 15. The reconfigurable load modulation amplifier of claim14 wherein the quadrature coupler load impedance is around about twice aterminal load impedance at the output load terminal.
 16. Thereconfigurable load modulation amplifier of claim 4 wherein a total loadimpedance presented to the carrier amplifier is relatively substantiallyhigher than a nominal saturated load impedance of the carrier amplifier.17. The reconfigurable load modulation amplifier of claim 4 wherein atotal load impedance presented to the carrier amplifier is relativelysubstantially lower than a nominal saturated load impedance of thecarrier amplifier.
 18. The reconfigurable load modulation amplifier ofclaim 4 wherein a total load impedance presented to the carrieramplifier is relatively substantially higher than a nominal saturatedload impedance of the reconfigurable load modulation amplifier at aroundfull power.
 19. The reconfigurable load modulation amplifier of claim 4wherein a total load impedance presented to the carrier amplifier isrelatively substantially lower than a nominal saturated load impedanceof load modulation amplifier at around full power.
 20. Thereconfigurable load modulation amplifier of claim 4 further includingvoltage standing wave ratio (VSWR) detection circuitry configured todetect a VSWR mismatch condition associated with the output loadterminal.
 21. The reconfigurable load modulation amplifier of claim 20wherein the VSWR detection circuitry is configured to determine if anantenna impedance is below or higher than a nominal output loadtermination impedance.
 22. The reconfigurable load modulation amplifierof claim 20 wherein the VSWR detection circuitry is configured to detectthat an antenna impedance is less than a nominal output load terminationimpedance, and then set an isolation impedance of the quadrature couplerto a termination impedance range that is greater than the nominal outputload termination impedance to improve isolation between the carrieramplifier and peak amplifier for improving load insensitive loadmodulation operation.
 23. The reconfigurable load modulation amplifierof claim 20 wherein the VSWR detection circuitry is configured to detectthat an antenna impedance is greater than a nominal output loadtermination impedance, and then set an isolation impedance of thequadrature coupler to a termination impedance range that is less thanthe nominal output load termination impedance to improve isolationbetween the carrier amplifier and the peak amplifier for improving loadinsensitive load modulation operation.
 24. The reconfigurable loadmodulation amplifier of claim 20 wherein the VSWR detection circuitry isconfigured to detect that an antenna impedance is less than a nominaloutput load termination impedance, and then set an isolation impedanceof the quadrature coupler to a termination impedance range that is lessthan the nominal output load termination impedance to improve isolationbetween the carrier amplifier and the peak amplifier for improving loadinsensitive inverted phase load modulation operation.
 25. Thereconfigurable load modulation amplifier of claim 20 wherein the VSWRdetection circuitry is configured to detect that an antenna impedance isgreater than a nominal output load termination impedance, and then setan isolation impedance of the quadrature coupler to a terminationimpedance range that is greater than the nominal output load terminationimpedance to improve isolation between the carrier amplifier and thepeak amplifier for improving load insensitive inverted phase loadmodulation operation.
 26. The reconfigurable load modulation amplifierof claim 20 wherein the VSWR detection circuitry is configured to detectphase and amplitude of a signal associated with power output through theoutput load terminal.
 27. The reconfigurable load modulation amplifierof claim 1 wherein the carrier amplifier is configured to operate from adrain voltage modulation.
 28. The reconfigurable load modulationamplifier of claim 1 wherein the peak amplifier is configured to operatefrom a drain voltage modulation.
 29. The reconfigurable load modulationamplifier of claim 1 where the carrier amplifier and the peak amplifierare both configured to operate from a drain voltage modulationsimultaneously.
 30. The reconfigurable load modulation amplifier ofclaim 1 wherein the quadrature coupler is a Lange coupler.